[SI-LIST] Re: Signal crossing Split plane
- From: Charles Harrington <ch_harrington@xxxxxxxxx>
- To: shlepnev@xxxxxxxxxxxxx, scott@xxxxxxxxxxxxx
- Date: Tue, 20 Nov 2007 14:45:54 -0800 (PST)
Yuriy,
I agree with some of your views. However, they contradict your via models.
I couldn?t reply yesterday, because I was trying search for the reference I
mentioned, since you needed it. Many other people replied off-line and so
needed the reference. Got it from IEEE Xplore.
A Novel Methodology for Defining the Boundaries of Geometrical
Discontinuities in Electronic Packages
Ndip, I.; Reichl, H.; Guttowski, S.;
Research in Microelectronics and Electronics 2006, Ph. D.
12- 15 June 2006 Page(s):193 - 196
You mentioned in your mail that the near field zone as a result of the
higher-order modes excited at the via expands with frequency and is very small.
I agree with you.
But the question is this. How small is it? How small or big is at 1 GHz, 10
GHz, 20 GHz? Have you ever studied it? You have to take this zone into
consideration when studying vias or any other structures that excite higher
order modes.
The method proposed in this paper is quite illustrative and useful. I
understand it this way (Please correct me if I understand it wrongly):
These higher-order modes (e.g., TE, TM...) are characteristics of the trace
or transmission line and they die exponentially away from the point of
excitation, i.e., the via-trace interface. S-parameters, like other network
parameters, give us the relation between input and output signals. Now, to
obtain S11, for example, you need to get the ratio of the reflected and input
signals. Both signals must be of the same "type". We can not directly compare
cars and aeroplanes, though both are used for transportation. You know your
input signal (e.g., a transverse electromagnetic wave), because you excited it
at the port. At discontinuities, an infinite order of given higher-order modes
can be excited. The orders or strength of the excited modes differ from one
discontinuity to another, although the modes can be the same. So, there is no
way you can know all the orders of the higher-order modes excited and how they
interact. Now if you place your ports quite close to the point
of excitation of these modes, then your S-parameters must be wrong. Why? In
this case, to obtain S11, you need to obtain the ratio of the unknown
higher-order modes and your known excited transverse electromagnetic wave at
the port. That?s why in most 3D full-wave solvers, it is recommended that ports
should be placed far away from the discontinuities, so as to enable these
higher-order modes to die. When they die, then you can easily define your
S-parameters which will then be the ratio of the input signal you know
(transverse electromagnetic wave) and the reflected signal you know (transverse
electromagnetic wave). To define the points where these modes die or have
attenuated substantially, these authors argued that near the discontinuity, the
imaginary part of the Poynting vector describes the reactive energy associated
with these higher-order modes. So they studied this imaginary part and used it
to define the point where the modes die. I think they mentioned that only
at a distance of about 1mm away from the via-trace interface, at 20 GHz (or
may be 30 GHz) may you place your ports, to get correct results. Certainly,
this depends on the via geometry and trace type. But I find the results very
helpful and can be used as a base for further experiments. You can get the
details from the paper.
Unfortunately in your case, you compare what you don?t know (reflected
signal) and what you know (excited input signal). In your via models, neither
did you define the required distance away from the via-trace interface needed
for these modes to die nor did you follow the advice given in full-wave solvers
to be far way from the via-trace interface. You considered the via just as the
barrel and the pads at 20 GHz and beyond. That?s why I mentioned yesterday that
your via models are not correct and your S-parameter results are misleading. If
you wish to study only the behaivor of the barrel alone at lower frequencies
(for what ever reason - but not for realistic designs), then you don't even
need a field solver. You can get formulas from good SI texts like that of
Horward Johnson or from papers.
At first I was also making the same mistakes as you are making right now. I
had a lot of difficulties to correlate my simulation and measurement results.
So I learnt a lot from this paper, from Professor C. Balanis (Advanced
engineering electromagnetics) and from Professor R. Collins (Field theory of
guided waves). I think these references will be good for you. You need all
three of them.
There are also a lot of points that you need to modify in your models.
It?s ridiculous when you talk of -30 dB attenuation of higher-order modes.
Which higher-order mode? Which order of this mode? Basic electromagnetic theory
teaches us that an infinite order of a given higher-order mode can be excited
at any discontinuity. An interaction between makes matters worst. So how do you
separate the different orders of the modes and tell which one attenuates by -30
dB? Are the modes propagating or evanescent? Never use rule of thumbs that have
no base. I supposed you meant attenuation of the fundamental mode which is
propagating.
I don?t know anything about the lumped ports you use. All I know is that some
lumped ports in some field solvers assume perfect H boundary conditions on the
sides. Consequently, depending you may not even capture stray fields. So you
can even get the worst results with lumped ports.
You can only shift your reference S-parameters plane and get accurate results
if your model captured all the necessary field behavior. But you can not
simulate the via and traces differently and then do some post-processing or
circuit modeling afterwards and expect to get correct results at higher
frequencies. The traces too are part of the ?via effect? at least, at the
frequencies you are interested in (20 GHz and beyond), because the stored
higher-order modes give rise to additional inductances and capacitances. These
inductances and capacitances can not be captured if you analyze the vias
separately from their traces.
Finally, the theory of multi-modal decomposition means different things to
different electrical engineers. So I don?t know what you mean. If you mean that
different parts of a system can be analyzed separately and then put together,
then it?s true that it has been done for decades now. But the question is this.
How do you bring the different parts together in the case where there are
discontinuities like vias? How do you define the via? How small or big is your
near field zone? I bet you, we have not yet understood this type of
decomposition and it has not been done, or at least published for decades.
Whenever we have to deal with vias and other discontinuities at higher
frequencies, straight-forward modeling can not be used.
Please Yuryi, don?t get me wrong. I?m not trying to highlight on your
errors. I have mine too, like any body else. No one is perfect. I?m just trying
to raise the point that we need to be careful when modeling vias at your
frequencies. I agree with most of the points you made, but disagree on the ones
stated above. We learn from each other when we exchange ideas about such
fundamental issues that affect our modeling results. I think that is one of the
reasons why Ray and his team set up this forum.
Best regards.
Charles
Yuriy Shlepnev <shlepnev@xxxxxxxxxxxxx> wrote: Charles,
I am sorry that the simulation examples were not helpful to you. I will
appreciate if you send me the reference you mentioned - I am preparing to be
shocked:)
You are absolutely right, the via-holes are not just pads and barrels and
there is no one solution that covers all possible cases. Analysis of
different vias has to be done in different ways. Transition to the traces
have to be almost always included in the final model for analysis of
multi-gigabit channels. Moreover sometime the via-hole problem cannot be
solved locally and require analysis of parallel plane structures with all
decoupling structures attached (see technical presentation #1 at
http://www.simberian.com/Presentations.php for more details on different
structures).
Considering the ports and excitation. Analysis of via-holes with lumped
ports provides just rough idea about the via-hole behavior. It is similar to
what you would see from a differential probe attached to the pads of the
via-holes. Transition to traces and transmission line or wave-ports have to
be used for the final extraction of S-parameters for the system-level
analysis (I am sorry that you missed this part in app notes). Note that it
is possible only for the localizable via-holes or via-holes not coupled to
parallel planes in general. Such t-line ports have to be positioned at a
distance from the via-hole that guaranties that the high-order modes are
attenuated substantially (for practical applications we usually use -30 dB
threshold at the highest frequency of interest). After such analysis, the
phase reference planes of S-parameters can be safely shifted closer to the
via-hole at the position where t-lines are still continuous to preserve
causality (to the edges of anti-pads for instance). Such transformation does
not affect the near field or high order modes around the via-holes and the
final model can be safely connected with the transmission line segments in a
system-level solver. Though, the model have to be used with transmission
line segments with length not less than in the electromagnetic analysis (to
avoid the near-field interaction between the vias and possible
discontinuities). This technique called the multi-modal de-compositional
analysis and used in microwave engineering for decades at frequencies even
higher than 20 GHz.
Note, that in typical PCB trace the cut-off frequencies for high-order modes
are extremely high. 10 mil trace on 10 mil dielectric with dielectric
constant 4.2 have cut-off frequency about 120 GHz, and the cross-over with
the surface TM mode may happen only at 200 GHz. Before 120 GHz the
high-order modes are evanescent and essentially form the via-hole near
field. This near-field zone is expanding with the frequency, but at 20 GHz
the area is still relatively small. Thus S-parameters only for the dominant
modes can be safely extracted and used as the via-hole model.
Cases when via-hole excite the non-evanescent parallel-plane modes and
planes are not stitched close to the via-hole cannot be solved locally (non
localizable) and may require the system-level analysis with all decoupling
structures attached.
Best regards,
Yuriy
Yuriy Shlepnev
Simberian Inc.
www.simberian.com
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx [mailto:si-list-bounce@xxxxxxxxxxxxx] On
Behalf Of Charles Harrington
Sent: Monday, November 19, 2007 8:33 AM
To: shlepnev@xxxxxxxxxxxxx; scott@xxxxxxxxxxxxx
Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST'
Subject: [SI-LIST] Re: Signal crossing Split plane
Yuriy,
not only are your slot simulations on your page not so helpful, but your
via simulations are very misleading. I think you'll run into trouble when
you try to compare your simulation and measurement results, because your
simulation models are unrealistic.
At such frequencies (20 GHz and beyond), the via can no longer be considered
to be just the barrel and the pads, as you did. The modes excited at the
via-trace interface don't die abruptly, but extend along the traces to the
ports. So either you seperate these modes from the originally excited modes
at the port (in order to obtain "clean" S-parameters') or you allow the
modes to die before they reach the ports (as recommended in most 3D
full-wave solvers).
I just read a very interesting research paper the other day on defining the
boundaries of discontinuties, in which these issues are properly examined. I
can't really remember the exact title nor its authors at the moment, but the
paper was presented at a Ph.D. research conference on microelectronics and
electronics somewhere in Europe (Italy, I presume). You'll be shocked at the
error you are making when you read this work.
You also connected the models of the via and transmission lines after the
simulations, correct? Here you go wrong again, because how do you know where
the vias "actually" begin and end? And at what freqency? These are very
complicated issues and I suggest you spend a little more time studying them
well.
Thanks.
Charles
Yuriy Shlepnev wrote: Scott,
I agree with you. It was just an illustration of a slot-type discontinuity
in general for some stackup configurations. It shows how a slot-type
discontinuity in a reference plane may reflect the signal even in the case
if slot does not cut across the board or around a patch (though, it might be
obvious for you). As soon as the coupling to a slot is strong, it has to be
simulated at the system level with a complete geometry of the slot or split,
with all relevant traces crossing the slot and all de-caps (if any). I
prefer to do it with the hybrid de-compositional approach on the base of
localized models built with an electromagnetic solver. The localized strip
to slot coupling effect can be captured with a 4-port S-parameter model for
strip crossing the slot for instance (two ports for the strip and two for
the slot). Combined with the strip and slot line models, it produces a
simple and computationally efficient system-level model that captures
practically all coupling and resonance effects.
Best regards,
Yuriy
Yuriy Shlepnev
Simberian Inc.
www.simberian.com
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx [mailto:si-list-bounce@xxxxxxxxxxxxx] On
Behalf Of Scott McMorrow
Sent: Sunday, November 18, 2007 12:29 PM
To: shlepnev@xxxxxxxxxxxxx
Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST'
Subject: [SI-LIST] Re: Signal crossing Split plane
Yuriy
Actually, these sorts of slot simulations are pretty meaningless. Slots
normally occur due to plane splits. As a result, the either extend from
one edge of a board to another edge, or when the plane is a square patch
the slot is a closed loop around the periphery of the plane. When this
happens, it is quite interesting to simulate multiple signals crossing
the slot. There is a very nice slot resonance mode that occurs that is
generally in the signal bandwidth (or at least 3rd harmonic) because of
the length of the slot. This induces a signficant amount of ringing and
crosstalk into neighboring traces.
scott
Scott McMorrow
Teraspeed Consulting Group LLC
121 North River Drive
Narragansett, RI 02882
(401) 284-1827 Business
(401) 284-1840 Fax
http://www.teraspeed.com
TeraspeedR is the registered service mark of
Teraspeed Consulting Group LLC
Yuriy Shlepnev wrote:
> Sunil,
>
> A simple example of how an electromagnetic solver can be used to
investigate
> the effect of a slot or split in a reference plane is provided at
> http://www.simberian.com/AppNotes.php - see the topmost app note.
>
> Best regards,
> Yuriy
>
> Yuriy Shlepnev
> Simberian Inc.
> www.simberian.com
>
> -----Original Message-----
> From: si-list-bounce@xxxxxxxxxxxxx [mailto:si-list-bounce@xxxxxxxxxxxxx]
On
> Behalf Of sunil bharadwaz
> Sent: Sunday, November 18, 2007 1:26 AM
> To: SI LIST
> Subject: [SI-LIST] Signal crossing Split plane
>
> Hi ,
> I have few signals (@ 80 Mhz & 20 Mhz) crossing the split Power
> plane in the adjacent layer.
>
> The 20 Mhz signal is diffrerential signal.The 80 Mhz is a single
> ended signal.
>
> I want to analyse the affect on Signal Integrity of these two
> signals due to split plane.
>
> I believe one need to define his stack up (Including the
> split) & then extract the layout to simulate.
>
> I'am not too sure if the prevalent SI tools have an option
> of creating split planes .
>
> Pls suggest me a right tool to carry out this.Also , i'am
> looking for a free tool to start with (even if the accuracy
> is slightly limited).
>
> Thanks in Advance!!
>
> Regards
> Sunil.Bh
>
>
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