Charles, You said:=20 " if you place your stitching vias (or whatever you call them), close to = the signal via, they will also define the inductance and impedance of the= return path. The inductance of the complete arrangment (signal via, stic= hing vias/return path) depends almost entirely on the geometry of the via= s and the distance between the stitching vias and signal via. If you incr= ease your simulation area without increasing the geometry/position of the= stitching vias, then the inductance/impedance of your return path will n= ot change." This is only partially true, but it is not the complete story. Dr Johnso= n is only partially correct on his treatment of via inductance. It is go= od as a 1st approximation, but it does not tell the whole story. His tre= atment assumes planar cavities of infinite extent and no full-wave effect= s occurring between the signal, stitch vias, and planes. Planes are not = infinite, and full-wave effects do occur, however, under some conditions = they can be approximated by the quaint TEM phenomena that we call "induct= ance". Any time vias penetrate a planar cavity, a circular parallel plat= e excitation will occur. Vias used to stitch the cavity cannot contain a= ll of the magnetic field and thus some energy propagates into the cavity,= depending on the frequency. The percentage of energy lost to the cavity= is frequency dependent, and will resonate and eventually be converted to= crosstalk, power system noise, and distortions in the signal as the ener= gy is re injected in the signal via. These effects are non-localizable w= ithout considering the complete planar boundary conditions, and constitut= e errors in modeling that are dependent on the exact configuration of the= planes. Many times these errors are acceptable. Some times they are not= =2E Frequency of operation, desired bandwidth, and required isolation ar= e determining factors in the acceptability of a particular modeling appro= ach. Placement of model launch ports at well-defined signal mode boundaries he= lps to solve some of the localization problems, but not all of them. Sig= nals following vias go through very tricky rabbit holes indeed, and as so= on as via lengths approach significant fractions of the signal wavelength= , large amounts of energy can be converted to parallel plate mode excitat= ion, well before other higher order modes of propagation develop. Yuriy and I have had many discussions about these things going back 5 or = 6 years now. I'm quite sure he has an ample grasp on the RF Electromagne= tic phenomenon. I've also looked at his application notes and find that = the are fine for what they are, but in some cases they are quick and dirt= y. There's nothing wrong with that! They serve to show and establish so= me fundamental points. Knowing Yuriy, I know that he understands the inh= erent limitations of the way that he launched into some of the structures= , and I find his emails quite enlightening as he expands on his thoughts.= Regards, Scott Scott McMorrow Teraspeed Consulting Group LLC 121 North River Drive Narragansett, RI 02882 (401) 284-1827 Business (401) 284-1840 Fax http://www.teraspeed.com Teraspeed=AE is the registered service mark of Teraspeed Consulting Group LLC Charles Harrington wrote: > Yuriy, > =20 > if you place your stitching vias (or whatever you call them), close t= o the signal via, they will also define the inductance and impedance of t= he return path. The inductance of the complete arrangment (signal via, st= iching vias/return path) depends almost entirely on the geometry of the v= ias and the distance between the stitching vias and signal via. If you in= crease your simulation area without increasing the geometry/position of t= he stitching vias, then the inductance/impedance of your return path will= not change. So your explanation that changing the simulation area automa= tically changes the impedance of the return path and |S11|, is not correc= t. If you do not believe me, then contact Dr. Howard Johnson for further = explanations. In his book (advanced black magic) he has an excellent expl= anation of the relationship between these stitching vias and the inductan= ce of the return current of the signal via. This is the only text I know = that treats these issues in a very > practical manner and I think it will be of great help to you. I strong= ly recommend it to every other person working with via modeling. Note tha= t the number of stitching vias you use depends on a number of other facto= rs. From his book, you'll see just a method. You can apply the method to = your own designs. Don't expect an automatic solution to your problem! > Remember, Maxwell never said Ampere is wrong. He said Ampere's circui= tal law is not complete and then completed it using the displacement curr= ent term (See J.C. Maxwell - On Physical Lines of Force, 1861). Now, when= your stitching vias define the return current path, the conduction curre= nt is much more greater than the inter-plane displacement currents, such = that we can use the original form of Ampere's law without any loss of acc= uracy. Quasi-static methods use this approximation too. It's not that the= displacement current is not present. But it is too small to be considere= d. With that said, you can visualize the discontinuity (considering its r= eturn paths and traces) as consisting of 2 parts. The closest part (which= you beautifully describe as the "near field zone") and the "far-field zo= ne" - if you allow me to use your words. In the "near field zone", which = should be less than 3mm for frequencies up to 50 GHz (from my experience)= when using typical PCB/package > geometries, |S11| (at a given constant frequency) depends almost compl= etely on the higher order modes excited by the discontinuity. |S11| decre= ases as the length increases because the higher order modes die off. In t= he far-field zone, |S11| increases because the higher order modes have co= mpeltely decayed and other factors now play the deciding role. As you kno= w, |S11| for a given line length at a givn frequency is constant.=20 > This is the method I have been using to define the boundaries of both= vertical discontinuties(such as via-holes) and horizontal discontinuties= (such as bends). And I'm fine with it. I have done many simulations and = measurements. As I said earlier, I got it from the paper I cited on Nov. = 20. and other publications from these authors. I had problems to accurate= ly model discontinuties at higher frequencies (up to 77 GHz) years ago. R= emember it is just a method and you have to take into consideration your = real design enviroment, if you apply the method proposed in the paper. Do= n't expect an automatic solution to your problem! As far I know, the term= "boundaries of discontinuities" as used in PCB/package design was first = introduced by these authors. If you have your own method, then you can al= so use it. But one thing is clear. You have to define these boundaries at= your 20 GHz and beyond. If not, your models are not correct and can not = be validated with measurements. I'm not > interested in your tool. Just like Chris, I'm interested in methods or= methodology as he calls it. > =20 > Hope this helps > =20 > Best regards > Charles > =20 > Yuriy Shlepnev <shlepnev@xxxxxxxxxxxxx> wrote: > Chris, > > I think all definitions of the interconnect models you provided may be > applicable under different circumstances. It depends on possibility to > localize discontinuities such as vias and splits for the electromagneti= c > analysis. If all discontinuities in you channel can be isolated for the= > electromagnetic analysis, then the de-compositional analysis you descri= bed > in a) and b) can be safely used. If such localization is impossible, th= e > system-level model has to be built either on the base of a complete 3D > electromagnetic analysis of the whole board (possible but not practical= ) or, > alternatively, the de-compositional model has to be extended with model= s of > such structures as parallel planes and splits with all decoupling struc= tures > connected to them (hybrid system-level models with transmission planes)= =2E > Those structures are the major reasons of non-localizability of > discontinuities on PCB and in packaging applications. > > How to define the boundaries of a discontinuity and localizability. If > simulation results (S-parameters for instance) are relatively independe= nt on > the simulation area size and on the boundary conditions, the discontinu= ity > can be isolated for the analysis and a reusable model can be generated > (otherwise the discontinuity is not localizable). Simple rules based on= line > width/substrate height can be used to define the simulation area in cas= e of > localizable discontinuities. Phase reference planes may be shifted towa= rd > the discontinuity to make it electrically smaller (for better fitting o= r > interpolation). A line segment with a minimal length to prohibit intera= ction > through the higher order modes have to be added at the system-level in = that > case. This de-compositional technique used in microwave engineering sin= ce > 40-s (Levin, Advanced theory of waveguides, 1951) and is the mainstream= in > the microwave system-level analysis tools. The smaller the localizable > discontinuity the smaller the effective discontinuity area. It provides= good > models even for micron-sized structures up to sub-mm wave frequencies. > > Note that dependency of S-parameters on the simulation area sometime ha= s > nothing to do with the higher order modes discussed here before, but ra= ther > related to parallel planes and to localizability. S-parameters of a sin= gle > via without or with a stitching via nearby can show significant depende= ncy > from the simulation area simply because of the discontinuity is not > completely localized and the impedance of a cavity formed by parallel p= lanes > may change the |S11| for instance. Increasing the simulation area, one > increase the inductance of the return path and together with the via > capacitance to the planes it may be visible as the decrease of |S11| wi= th > the increase of the simulation area size (effect observed in the paper = cited > below). Such problems may be on the border line between the localizable= and > non-localizable problems and sometime may even require the system-level= > hybrid models with the transmission planes. > > Who has to define the boundaries of discontinuities or minimal length o= f the > line segments to connect the discontinuities. Ideally, it has to be the= > system-level tool that decomposes a channel into transmission lines, > discontinuities and possibly transmission planes. All mainstream SI too= ls > are already based on the decomposition into line segments. It may inclu= de > coupling between the lines bases on physical or electrical thresholds. = The > same approach has to be used to define what discontinuities may be anal= yzed > with a 3D EM tool and what discontinuities require hybrid models. If tw= o > discontinuities are too close to each other (physical or electrical cri= teria > can be used) - they have to be analyzed in a 3D solver as a whole and s= o on. > In addition, a 3D solver has to define sufficient simulation area > automatically and produce the model that is electrically as small as > possible. Without such interaction between the system-level tool and a = 3D > solver you have to follow the recommendations provided by a 3D tool ven= dor > and make sure that the discontinuity model is connected in the final de= sign > with sufficient line segments. I think that report on the minimal lengt= h of > the line segments would be a good feature for an electromagnetic tool. > > Best regards, > Yuriy > > Yuriy Shlepnev, > Simberian Inc. > www.simberian.com > > -----Original Message----- > From: Chris Cheng [mailto:Chris.Cheng@xxxxxxxx]=20 > Sent: Thursday, November 29, 2007 6:18 PM > To: ch_harrington@xxxxxxxxx; shlepnev@xxxxxxxxxxxxx > Cc: SI LIST > Subject: RE: [SI-LIST] Re: Signal crossing Split plane > > Charles and Yuriy, > I have a philosophical question about modeling these 3D structures. > It seems both of you agree that the entry ports needs to be back out to= > certain distant from the structure itself (most likely dimensionally > compatible to the structure itself). So what is the definition of the > overall system level interconnect model ? > One can have the following defintions : > a) interconnect model (most likely lossy trace model) with length up to= the > extended port location + the 3-D model of the plane cut/via transition = model > b) interconnect model (most likely lossy trace model) as report by the > design data base + the 3-D model of the plane cut/via transition model = - > effect of just the extend port length of the interconnect model > c) the entire interconnect structure is simulation in one gigantic 3D > structure=20 > > The combine last two terms of b) is what Roger Harrington used to call > excess parasitics.=20 > In a PCB interconnect environment, a) and b) for all practical purpose = are > the same because the interconnect length >> extended port length. But f= or > package model where the entire structures are measured in mm or mils, a= ) and > b) > may have significant differences.=20 > > Should a 3D cad tool report the "excess parasitics" so that users can s= imply > use the length report of the design database, then add in the via/plan = cut > section anytime he/she encounters such structure ? > Or should a 3D cad tool be just modeling the true 3D structure but then= has > to warn users to back out the interconnect trace length to account for = the > extend port length (which seems to require careful consideration of the= 3D > structure on a case by case basis). > Or, just lump the 3D structure into a gigantic 3D file together with th= e > rest of the interconnect and pray that the simulator will converge ? > > > -----Original Message----- > From: si-list-bounce@xxxxxxxxxxxxx > [mailto:si-list-bounce@xxxxxxxxxxxxx]On Behalf Of Charles Harrington > Sent: Wednesday, November 21, 2007 2:57 PM > To: shlepnev@xxxxxxxxxxxxx > Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST' > Subject: [SI-LIST] Re: Signal crossing Split plane > > > > Yuriy, > I think we really have to end the discussion. I recommend you also > talk to some experts in this forum about your models. They will tell yo= u > exactly what I=12m trying to say and even more. > I didn=12t even know you have your own software. But I cannot understan= d why > you make such claims that your software can compute =13whatever multila= yered > geometry=14 and that =13it also automatically defines the boundary of t= he > discontinuities=14. You know this is not true. We all know this is not = true. > So why do you make such claims? If I ask you, with what degree of accur= acy > does your software compute "whatever multilayer geometry" (when compare= d > to measurements) and how does it automatically define the boundaries of= > discontinuities, I know that you will be baffled. So, I don=12t need th= e > answers to these questions. However, I'm glad you acknowledge the fact > that you need about 1mm distance away from the via pad at such higher > frequencies to get accurate results. Let us leave it there. I will not > write any more. > I wish you the best with your models. 25 yrs of experience is quite a l= ot.. > I respect that. But as you can see, there is still a lot out there to > learn. > Best regards > Charles > > > > Yuriy Shlepnev wrote: Hi Charles, > > Thank you for the reference. I am familiar with this paper as well as w= ith > the other publications of this group from Fraunhofer Institute. > First of all, our 3D full-wave solver allows to build different via-hol= e > models. It solves whatever multilayered geometry with ports you put in > there. > Second, the solver automatically defines the boundary of the > discontinuities. See for instance the final model for optimal via-hole = on > slide 8 in > http://www.simberian.com/Papers/OptimalDifViaholesDesign6pPCB.pdf. The > differential line segment length in that particular example is about 1 = mm, > that is sufficient for high-order modes to die even at 30 GHz. Though, = to > define the area we use technique different from one described in the pa= per > (I hinted details earlier). Lumped ports are often used for the > preliminary > optimization of via-holes because of it is quick and it provides good > approximation (see for instance the final model and comments in the > presentation mentioned above). Essentially, it ends the discussion. > If you looked through the app notes on our web page you just saw the ti= p > of > an iceberg. We put about 25 years of research to develop and validate t= he > technology. It is well documented on our web site in Downloads/Papers a= nd > Presentations areas. > And, I do not even want to start discuss the definition of ports or > multimodal decomposition, because of it looks strange to me that after > reading Collins you still do not understand what it means and how it > applies > to the multilayered circuits. > > Best regards, > Yuriy > > Yuriy Shlepnev, Ph.D. > President, Simberian Inc. > 2326 E Denny Way, Seattle, WA 98122, USA > Tel/fax +1-206-726-1098 > Cell +1-206-409-2368 > Skype shlepnev > > www.simberian.com > > > > > > From: Charles Harrington [mailto:ch_harrington@xxxxxxxxx] > Sent: Tuesday, November 20, 2007 2:46 PM > To: shlepnev@xxxxxxxxxxxxx; scott@xxxxxxxxxxxxx > Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST' > Subject: RE: [SI-LIST] Re: Signal crossing Split plane > > Yuriy, > I agree with some of your views. However, they contradict your via mode= ls.. > I couldn't reply yesterday, because I was trying search for the referen= ce > I > mentioned, since you needed it. Many other people replied off-line and = so > needed the reference. Got it from IEEE Xplore. > > A Novel Methodology for Defining the Boundaries of Geometrical > Discontinuities in Electronic Packages > Ndip, I.; Reichl, H.; Guttowski, S.; > Research in Microelectronics and Electronics 2006, Ph. D. > 12- 15 June 2006 Page(s):193 - 196 > > You mentioned in your mail that the near field zone as a result of the > higher-order modes excited at the via expands with frequency and is ver= y > small. I agree with you. > But the question is this. How small is it? How small or big is at 1 GHz= , > 10 > GHz, 20 GHz? Have you ever studied it? You have to take this zone into > consideration when studying vias or any other structures that excite > higher > order modes. > The method proposed in this paper is quite illustrative and useful. I > understand it this way (Please correct me if I understand it wrongly): > These higher-order modes (e.g., TE, TM...) are characteristics of the > trace > or transmission line and they die exponentially away from the point of > excitation, i.e., the via-trace interface. S-parameters, like other > network > parameters, give us the relation between input and output signals. Now,= to > obtain S11, for example, you need to get the ratio of the reflected and= > input signals. Both signals must be of the same "type". We can not > directly > compare cars and aeroplanes, though both are used for transportation. Y= ou > know your input signal (e.g., a transverse electromagnetic wave), becau= se > you excited it at the port. At discontinuities, an infinite order of > given > higher-order modes can be excited. The orders or strength of the excite= d > modes differ from one discontinuity to another, although the modes can = be > the same. So, there is no way you can know all the orders of the > higher-order modes excited and how they interact. Now if you place your= > ports quite close to the point of excitation of these modes, then your > S-parameters must be wrong. Why? In this case, to obtain S11, you need = to > obtain the ratio of the unknown higher-order modes and your known excit= ed > transverse electromagnetic wave at the port. That's why in most 3D > full-wave > solvers, it is recommended that ports should be placed far away from th= e > discontinuities, so as to enable these higher-order modes to die. When > they > die, then you can easily define your S-parameters which will then be th= e > ratio of the input signal you know (transverse electromagnetic wave) an= d > the > reflected signal you know (transverse electromagnetic wave). To define = the > points where these modes die or have attenuated substantially, these > authors > argued that near the discontinuity, the imaginary part of the Poynting > vector describes the reactive energy associated with these higher-order= > modes. So they studied this imaginary part and used it to define the po= int > where the modes die. I think they mentioned that only at a distance of > about > 1mm away from the via-trace interface, at 20 GHz (or may be 30 GHz) may= > you > place your ports, to get correct results. Certainly, this depends on th= e > via > geometry and trace type. But I find the results very helpful and can be= > used > as a base for further experiments. You can get the details from the pap= er.. > Unfortunately in your case, you compare what you don't know (reflected > signal) and what you know (excited input signal). In your via models, > neither did you define the required distance away from the via-trace > interface needed for these modes to die nor did you follow the advice > given > in full-wave solvers to be far way from the via-trace interface. You > considered the via just as the barrel and the pads at 20 GHz and beyond= =2E > That's why I mentioned yesterday that your via models are not correct a= nd > your S-parameter results are misleading. If you wish to study only the > behaivor of the barrel alone at lower frequencies (for what ever reason= - > but not for realistic designs), then you don't even need a field solver= =2E > You > can get formulas from good SI texts like that of Horward Johnson or fro= m > papers. > At first I was also making the same mistakes as you are making right no= w. > I > had a lot of difficulties to correlate my simulation and measurement > results. So I learnt a lot from this paper, from Professor C. Balanis > (Advanced engineering electromagnetics) and from Professor R. Collins > (Field > theory of guided waves). I think these references will be good for you.= > You > need all three of them. > There are also a lot of points that you need to modify in your models. > It's ridiculous when you talk of -30 dB attenuation of higher-order mod= es.. > Which higher-order mode? Which order of this mode? Basic electromagneti= c > theory teaches us that an infinite order of a given higher-order mode c= an > be > excited at any discontinuity. An interaction between makes matters wors= t. > So > how do you separate the different orders of the modes and tell which on= e > attenuates by -30 dB? Are the modes propagating or evanescent? Never us= e > rule of thumbs that have no base. I supposed you meant attenuation of t= he > fundamental mode which is propagating. > I don't know anything about the lumped ports you use. All I know is tha= t > some lumped ports in some field solvers assume perfect H boundary > conditions > on the sides. Consequently, depending you may not even capture stray > fields. > So you can even get the worst results with lumped ports. > You can only shift your reference S-parameters plane and get accurate > results if your model captured all the necessary field behavior. But yo= u > can > not simulate the via and traces differently and then do some > post-processing > or circuit modeling afterwards and expect to get correct results at hig= her > frequencies. The traces too are part of the "via effect" at least, at t= he > frequencies you are interested in (20 GHz and beyond), because the stor= ed > higher-order modes give rise to additional inductances and capacitances= =2E > These inductances and capacitances can not be captured if you analyze t= he > vias separately from their traces. > Finally, the theory of multi-modal decomposition means different things= to > different electrical engineers. So I don't know what you mean. If you m= ean > that different parts of a system can be analyzed separately and then pu= t > together, then it's true that it has been done for decades now. But the= > question is this. How do you bring the different parts together in the > case > where there are discontinuities like vias? How do you define the via? H= ow > small or big is your near field zone? I bet you, we have not yet > understood > this type of decomposition and it has not been done, or at least publis= hed > for decades. Whenever we have to deal with vias and other discontinuiti= es > at > higher frequencies, straight-forward modeling can not be used. > Please Yuryi, don't get me wrong. I'm not trying to highlight on your > errors. I have mine too, like any body else. No one is perfect. I'm jus= t > trying to raise the point that we need to be careful when modeling vias= at > your frequencies. I agree with most of the points you made, but disagre= e > on > the ones stated above. We learn from each other when we exchange ideas > about > such fundamental issues that affect our modeling results. I think that = is > one of the reasons why Ray and his team set up this forum. > > Best regards. > Charles > > > > > > > > > Yuriy Shlepnev wrote: > Charles, > > I am sorry that the simulation examples were not helpful to you. I will= > appreciate if you send me the reference you mentioned - I am preparing = to > be > shocked:) > > You are absolutely right, the via-holes are not just pads and barrels a= nd > there is no one solution that covers all possible cases. Analysis of > different vias has to be done in different ways. Transition to the trac= es > have to be almost always included in the final model for analysis of > multi-gigabit channels. Moreover sometime the via-hole problem cannot b= e > solved locally and require analysis of parallel plane structures with a= ll > decoupling structures attached (see technical presentation #1 at > http://www.simberian.com/Presentations.php for more details on differen= t > structures). > > Considering the ports and excitation. Analysis of via-holes with lumped= > ports provides just rough idea about the via-hole behavior. It is simil= ar > to > what you would see from a differential probe attached to the pads of th= e > via-holes. Transition to traces and transmission line or wave-ports hav= e > to > be used for the final extraction of S-parameters for the system-level > analysis (I am sorry that you missed this part in app notes). Note that= it > is possible only for the localizable via-holes or via-holes not coupled= to > parallel planes in general. Such t-line ports have to be positioned at = a > distance from the via-hole that guaranties that the high-order modes ar= e > attenuated substantially (for practical applications we usually use -30= dB > threshold at the highest frequency of interest). After such analysis, t= he > phase reference planes of S-parameters can be safely shifted closer to = the > via-hole at the position where t-lines are still continuous to preserve= > causality (to the edges of anti-pads for instance). Such transformation= > does > not affect the near field or high order modes around the via-holes and = the > final model can be safely connected with the transmission line segments= in > a > system-level solver. Though, the model have to be used with transmissio= n > line segments with length not less than in the electromagnetic analysis= > (to > avoid the near-field interaction between the vias and possible > discontinuities). This technique called the multi-modal de-compositiona= l > analysis and used in microwave engineering for decades at frequencies e= ven > higher than 20 GHz. > Note, that in typical PCB trace the cut-off frequencies for high-order > modes > are extremely high. 10 mil trace on 10 mil dielectric with dielectric > constant 4.2 have cut-off frequency about 120 GHz, and the cross-over w= ith > the surface TM mode may happen only at 200 GHz. Before 120 GHz the > high-order modes are evanescent and essentially form the via-hole near > field. This near-field zone is expanding with the frequency, but at 20 = GHz > the area is still relatively small. Thus S-parameters only for the > dominant > modes can be safely extracted and used as the via-hole model. > Cases when via-hole excite the non-evanescent parallel-plane modes and > planes are not stitched close to the via-hole cannot be solved locally > > =3D=3D=3D message truncated =3D=3D=3D > > =20 > --------------------------------- > Be a better friend, newshound, and know-it-all with Yahoo! Mobile. Try= it now. > > ------------------------------------------------------------------ > To unsubscribe from si-list: > si-list-request@xxxxxxxxxxxxx with 'unsubscribe' in the Subject field > > or to administer your membership from a web page, go to: > //www.freelists.org/webpage/si-list > > For help: > si-list-request@xxxxxxxxxxxxx with 'help' in the Subject field > > > List technical documents are available at: > http://www.si-list.net > > List archives are viewable at: =20 > //www.freelists.org/archives/si-list > or at our remote archives: > http://groups.yahoo.com/group/si-list/messages > Old (prior to June 6, 2001) list archives are viewable at: > http://www.qsl.net/wb6tpu > =20 > > > =20 ------------------------------------------------------------------ To unsubscribe from si-list: si-list-request@xxxxxxxxxxxxx with 'unsubscribe' in the Subject field or to administer your membership from a web page, go to: //www.freelists.org/webpage/si-list For help: si-list-request@xxxxxxxxxxxxx with 'help' in the Subject field List technical documents are available at: http://www.si-list.net List archives are viewable at: //www.freelists.org/archives/si-list or at our remote archives: http://groups.yahoo.com/group/si-list/messages Old (prior to June 6, 2001) list archives are viewable at: http://www.qsl.net/wb6tpu