[SI-LIST] Re: SSO and load capacitance
- From: Larry Smith <LSMITH@xxxxxxxxxx>
- To: 'Scott McMorrow' <scott@xxxxxxxxxxxxx>, "'si-list@xxxxxxxxxxxxx'" <si-list@xxxxxxxxxxxxx>
- Date: Tue, 2 Aug 2011 11:55:58 -0700
Scott - You are correct about the reflection phenomenon with transmission lines
that are not perfectly terminated. We will definitely get a bounce off the
load capacitance and energy coming back towards the place where SSN originated.
I consider that to be more of a bus termination problem than an SSN problem.
I have a very narrow definition of SSN (simultaneous switch noise). It is the
waveform launched into a victim (quiet-high or quiet-low) signal line when
aggressor signals switch simultaneously. As mentioned before, there is an
inductive coupling portion to the signal that is only launched during the
aggressor rise (fall) time and there is a longer wavelength PDN component (that
Steve referred to) that is initiated by the SSN event. The inductive coupling
component tends to be about 200 pSec long (~ .35/.2 = 1.75 GHz content) and the
PDN component tends to be at about 100MHz.
I consider the reflections that you have discussed to be part of the "system
response" to the SSN event. Yes, this is very important but I find it useful
to divide things up into the initial noise stimulus and then the system
response to that stimulus because you may choose to manage each of these
independently. As a component house, we can only manage the initial stimulus.
The system house needs to manage the board/reflection/termination/load issues.
And of course, we need to provide the system houses with components that they
can live with on their boards. :)
Another point to be made is that the inductive coupling glitch (1.75MHz
frequency content) fortunately suffers significant loss on its journey to the
load and back to the source. This tends to damp out system resonances at least
to some degree. When we go to measure inductive coupling SSN, we have to
account for the loss in our measurement fixture (possibly 18 inches of board
transmission line) and use de-embedding to find out the SSN amplitude that our
customers are likely to see at some distance away from the SSN source. In this
case, loss is our friend..
Regards,
Larry
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx [mailto:si-list-bounce@xxxxxxxxxxxxx] On
Behalf Of Scott McMorrow
Sent: Tuesday, August 02, 2011 9:17 AM
To: si-list@xxxxxxxxxxxxx
Subject: [SI-LIST] Re: SSO and load capacitance
Larry
Be careful. Load capacitance looks like a momentary reflected short
circuit to the driver. If the driver is not well matched (for example
low output impedance drive on a DDR driver), we have effectively two low
impedance discontinuities on either end of the line, setting up a
half-wave resonant circuit. Now, you are quite right that the reflected
discontinuity back from the load capacitance can never be larger than
the initial SSO charge up of the line at the driver, however, it is
possible to tune the delay such that the reflection reaches the driver
at exactly the wrong time - when the driver switches during the next
cycle. With the right bit rate, a standing wave can occur, causing SSO
to peak.
Scott
Scott McMorrow
Teraspeed Consulting Group LLC
121 North River Drive
Narragansett, RI 02882
(401) 284-1827 Business
(401) 284-1840 Fax
http://www.teraspeed.com
Teraspeed(r) is the registered service mark of
Teraspeed Consulting Group LLC
On 8/2/2011 11:18 AM, Larry Smith wrote:
> Jason - the comment is correct: the worst case SSN waveforms will be found
> with minimum load capacitance. But some explanation is required.
>
> First, SSN can be broken down into two components: inductive coupling and
> PDN. Steve is referring to the PDN part in his response. But usually the
> greatest SSN noise amplitude measured at the far end of a signal transmission
> line comes from inductive coupling, not PDN.
>
> Inductive coupling is related to mutual inductance between aggressor signals
> and the victim signal. It only happens during the rise (fall) time of the
> driver because that is when the di/dt takes place. To a first approximation,
> the voltage noise that gets launched into a victim transmission line (under
> the BGA that makes the SSN) is proportional to m*di/dt where m is the sum of
> the mutual inductance from all the aggressors to the victim and i is the
> current in the aggressors. Mutual inductance occurs in the wire bonds,
> package vias, balls and PCB vias and to a first approximation is proportional
> to the length of these structures.
>
> These days, the aggressor rise time is on the order of 200pSec, which is the
> time that it takes signals to travel about an inch down a transmission line.
> The capacitance load in question is down at the far end of the transmission
> line, let's assume 6 inches. The 200pSec rise time aggressors launch an SSN
> noise pulse into the victim signal net that is approximately 200pSec wide and
> it arrives at the capacitance load about 1000pSec later. The load
> capacitance at the far end will have no effect on the SSN event that launches
> the SSN glitch into the victim transmission line.
>
> When the SSN glitch arrives at the far end of the transmission line, it often
> finds a 50 ohm termination. The noise measured at the far end is identical
> to the glitch launched into the near end, assuming lossless lines. Now if
> there is any capacitance load at the far end, glitch energy goes into
> charging up the load. The measured SSN glitch voltage amplitude will be less
> with more load capacitance.
>
> Regards,
> Larry Smith
>
> -----Original Message-----
> From: si-list-bounce@xxxxxxxxxxxxx [mailto:si-list-bounce@xxxxxxxxxxxxx] On
> Behalf Of steve weir
> Sent: Tuesday, August 02, 2011 4:41 AM
> To: si-list@xxxxxxxxxxxxx
> Subject: [SI-LIST] Re: SSO and load capacitance
>
> Oops the second formula was energy not charge. It should have read
> Qload comes from Qbypass = Vdroop*Cbypass.
>
> Steve
> On 8/2/2011 4:24 AM, steve weir wrote:
>> Jason, there are two possible sources of confusion. The first is
>> possible confusion between output load capacitance with die capacitance
>> per output driver. Your intuition is correct: If we simplify the PDN /
>> driver network to a switched capacitor representation, then we deposit
>> Qload = Vdd*Cload on each output line that switches from low to high,
>> and remove Qload from each output that switches from high to low. For
>> the low to high switching outputs: Qload comes from Qbypass = (Vdd -
>> Vdroop)^2/2*Cbypass.
>>
>> The second source of confusion comes from the fact that any loads that
>> remain statically high can draw current from any load capacitance that
>> connects to the driver outputs, supporting other outputs that switch
>> from low to high.
>>
>> Steve.
>>
>>
>> On 8/2/2011 3:01 AM, Jason Young wrote:
>>> Dear Experts,
>>> I have read a couple of documents are from silicon IP vendors discussing
>>> the number of power/ground pads needed to meet SSO requirements for a given
>>> number of output drivers. These documents mention that worse case
>>> conditions for SSO are with the smallest output load capacitance. At first
>>> this seems counter intuitive. My initial reasoning would be that a larger
>>> capacitance would present a lower impedance load and hence greater dI/dt,
>>> greater IR drop and greater supply rail bounce. Could you please help me
>>> understand?
>>> Regards,
>>> Jason
>>>
>>>
>>>
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>
> --
> Steve Weir
> IPBLOX, LLC
> 150 N. Center St. #211
> Reno, NV 89501
> www.ipblox.com
>
> (775) 299-4236 Business
> (866) 675-4630 Toll-free
> (707) 780-1951 Fax
>
>
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