[SI-LIST] Re: Split planes and ground return wires
- From: "Dr. Howard Johnson" <howiej@xxxxxxxxxx>
- To: <si-list@xxxxxxxxxxxxx>
- Date: Thu, 20 Feb 2003 13:47:56 -0800
Dear Charles,
I'm going to address your question first with an excerpt
from
one of my previous articles, "Cable Shield Grounding".
In high-speed digital applications, a low
impedance
connection between the shield and the equipment chassis
*at
both ends* is required in order for the shield to do its
job.
The shield connection impedance must be low in the
frequency
range over which you propose for the shield to operate.
The
measure of shield connection efficacy for a
high-speed
connector is called the ground transfer impedance, or
shield
transfer impedance, of the connector, and it is a
crucial
parameter.
In low-speed applications involving high-impedance
circuitry,
where most of the near-field energy surrounding
the
conductors is in the electric field mode (as opposed to
the
magnetic field mode), shields need only be grounded at
one
end. In this case the shield acts as a Faraday
cage
surrounding the conductors, prevent the egress (or
ingress)
of electric fields.
In high-speed applications involving low-impedance
circuitry,
most of the near-field energy surrounding the
conductors is
in the magnetic field mode, and for that problem,
only a
magnetic shield will work. That's what the
double-grounded
shield provides. Grounding both ends of the shield
permits
high-frequency currents to circulate in the shield,
which
will counteract the currents flowing in the
signal
conductors. These counteracting currents create
magnetic
fields that cancel the magnetic fields emanating from
the
signal conductors, providing a magnetic shielding effect.
For the magnetic shield to operate properly, we must
provide
means for current to enter (or exit) at both ends of
the
cable. As a result, a low-impedance connection to
the
chassis, operative over the frequency range of our
digital
signals, is required that *both* ends of our shielded
cable.
(See Henry Ott, "Noise Reduction Techniques in
Electronic
Systems", 2nd ed., John Wiley & Sons, 1988.)
There are exceptions to the "ground coax at both ends rule".
These
exceptions involve cases where for some reason no currents
enter
or leave the end-terminal equipment. For example, suppose
that
at the receiving equipment the coaxial cable feeds the
primary
winding of an ideal transformer. Let the secondary winding
of the
transformer feed the receiver circuitry. There is thus no DC
connection (no common-mode current connection) through the
transformer. Of course, in actual practice you do get
current
flowing through the parasitic capacitance of the transformer
from primary to secondary, and this current then must make
its way
back to the source, which in your case is one of those big
grounding straps between the cabinets. In a high-speed
digital
product you would therefore likely not pass your radiated
emissions tests with that architecture.
Another approach useful at lower speeds is to terminate the
coaxial cable to itself with 50 ohms (or whatever is
appropriate)
and then feed only the signal wire into your receiver. A
suitably high-impedance receiver input draws very little
current,
thereby limiting the current that must flow back to the
source
on the big grounding straps. Once again, at high speeds the
current required to drive the parasitic capacitance of the
receiver is not insubstantial, and so this architecture
won't
work.
Your old application probably used some combination of
either
transformer coupling, a high-impedance receiver, or both.
(Common-mode chokes can also work).
One approach that works well at speeds up to 125 Mbaud on
category-5 unshielded twisted pairs is to use a well
balanced
transformer (sometimes in series with a common-mode choke)
to
limit the common-mode current flowing into the receiver (or
out
of the transmitter). Such architectures work and pass FCC/EN
radiated emissions tests at speeds up to 125 Mbaud with
output signals on the order of 1 V p-p.
Best regards,
Dr. Howard Johnson, Signal Consulting Inc.,
tel +1 509-997-0505, howiej@xxxxxxxxxx
http:\\sigcon.com -- High-Speed Digital Design articles,
books, tools, and seminars
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx
[mailto:si-list-bounce@xxxxxxxxxxxxx]On Behalf Of Charles
Sweeney
Sent: Thursday, February 20, 2003 10:36 AM
To: howiej@xxxxxxxxxx; si-list@xxxxxxxxxxxxx
Subject: [SI-LIST] Re: Split planes and ground return wires
Howard,
This makes me wonder about the situation of connecting
together pieces
of equipment each with their own separate power supply. I
used to design
radar systems composed of multiple racks of equipment, and
we were
concerned about ground loops. One technique we used was to
connect all
the grounds to a single point (a star configuration) using
very thick
earth braid, and then all the grounds in the cables (eg
shielded ribbon
cable) were connected to ground at only one end of each
cable. The
signals therefore had a reference ground plane very close,
but the
actual return path was a very long way away.
We could have connected the other cable ends to ground
through a
capacitor to provide the AC return path, but would this have
been a
"big" improvement? I suppose my question comes down to, is
it the close
presence of the ground plane that is important, or does the
ground plane
have to carry the return current.
This has implications in PCB's where striplines may have
power planes
(not ground) on both sides, but the ground plane carrying
the return
current could be several layers away. This is an
increasingly likely
scenario in PCB's with different chips having different VCCs
and so
needing several power planes.
Regards
Charles
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx
[mailto:si-list-bounce@xxxxxxxxxxxxx]
On Behalf Of Dr. Howard Johnson
Sent: 20 February 2003 17:37
To: si-list@xxxxxxxxxxxxx
Subject: [SI-LIST] Re: Split planes and ground return wires
Dear Marc,
You have raised what is (to me) a very interesting question,
one which deserves a good analytical answer.
If we just deal with a group of ten signals surrounding one
nearby ground return wire, there are some analytical tools
that can help you predict the answer.
First, you need to know the mutual inductance between
signals
in your configuration.
Given an aggressive wire at position a, and a victim wire at
position v, and one and only one ground wire at position g,
all traversing z-axis distance h, if I recall correctly the
formula for the mutual inductance shared between the
aggressor
and victim is:
Lm = 5.08 * h * [ln(|v-g|/R) - ln(|v-a|/|g-a|)]
where R is the radius of the wire,
the dimensions h, |v-g|, |v-a|, |g-a|, and R are inches,
the function ln() is the natural logarithm, and
the answer Lm appears in nH
As you can see the wire radius R appears as a logarithmic
term in the equaltion and so changes in the wire radius do
not strongly affect the outcome. The height h of the cavity
between the boards (the wire length) affects the result
directly and is thus your most powerful control variable.
Let's do an example with the positions a, v, and g in a
straight row with equal spacing s=0.1 inch.
|v-a| = .1
|v-g| = .1
|g-a| = .2
R = .01 (radius of 24 AWG wire)
h = .2 (from your note, although this sounds awfully
tight)
Lm = 5.08 * .2 * [ln(.1/.01) - ln(.1/.2)]
Lm = 3.04 nH
You should re-run this example assuming the only return
paths
are provided by the so-called "low inductance" ground pins
in your backplane connector. Because they are located
further
away from the signals, the mutual inductance Lm will come
out
larger -- thus the importance of locating the return paths
near the signals.
The next issue has to do with your signal risetimes. The
risetime (not the operating speed) determines the bandwidth
of
your signals. Assuming you are working with a modest CMOS
process having rise/fall times on the order of Tr = 1 nS,
the
bandwidth (upper-bounded by approximately 0.5/Tr) works out
to f = 500 MHz, at which frequency the impedance Zm
represented
by the mutual inductor Lm at frequency f
equals Zm = 2*pi*f*Lm = 9.55 ohms.
The crosstalk you receive has to do with the ratio of Zm to
the
circuit impedance Z0. If you were using end-terminated
transmission
lines with Z0=50 ohms, the numbers in this case indicate a
crosstalk of approximately 9.55/50 = 19 % FOR EACH
AGGRESSOR.
Fortunately, in your application the wires are quite short
compared
to 1 ns and so they are likely not terminated, in which case
the
current drawn on each wire (and thus the mutual-inductive
crosstalk)
will be considerably less than 19 %.
By the way, if you are using source-terminated lines the
current
waveform produced on the wire appears as two steps, one
positive
followed by one negative, with amplitudes equal to
(1/2)Vcc/Z0
and separation in time approximately equal to the line delay
(depending on where along this line your "connector"
appears).
If the line length, and thus the separation of the two
opposing
current waveforms, is small compared to the risetime the
two opposing chunks of mutual-inductive crosstalk created in
this configuration tend to cancel out.
Lastly, the crosstalk that concerns you in this example
bears a
striking resemblance to the crosstalk that occurs when
ordinary
vias traverse a pcb. In analyzing those situations one takes
into
account all the nearby return paths, which tends to diminish
the
value of Lm quoted above for a single return path. Software
for
such calculations is provided by, among other companies,
Sigrity.
Best regards,
Dr. Howard Johnson, Signal Consulting Inc.,
tel +1 509-997-0505, howiej@xxxxxxxxxx
http:\\sigcon.com -- High-Speed Digital Design articles,
books, tools, and seminars
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx
[mailto:si-list-bounce@xxxxxxxxxxxxx]On Behalf Of Godbout,
Marc
Sent: Wednesday, February 19, 2003 6:57 AM
To: si-list@xxxxxxxxxxxxx
Subject: [SI-LIST] Split planes and ground return wires
I've got questions concerning ground return wire
requirements and hope the
list can help. First the configuration.
Our system as got boards that are physically two separate
PCB's, about 6" X
7", sandwiching, and bonded to, a metal plate. Signals need
to traverse from
one PCB to another. To accomplish this, we run wires through
holes in the
plate. These signals are basically microprocessor-type
signals, 5V CMOS. The
bus speeds are 25MHz or lower.
Ground returns are added at about a 10:1 ratio, and placed
somewhat near the
groups of signals. The ground planes are also connected to
each other
through the backplane connector through 40 low-inductance
pins.
So I see this as a not-quite classical split plane problem.
The split is
crossed by 40 connector wires and maybe 15 z-axis wires,
each 0.200" long.
We've got some mechanical constraints that are making me
look into the
minimum requirements of the ground return wires. I feel that
the "bigger is
better" theory was used to design them in the first place,
but I've got to
cut back on that right now. So the question is, by how much?
So what are the areas or major concerns that I should
investigate? How do I
go about calculating a minimum wire size? And are there any
texts that would
help me zero in on my analysis methods?
Thanks for any help you could give.
-Marc
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