[SI-LIST] Re: Signal crossing Split plane
- From: Charles Harrington <ch_harrington@xxxxxxxxx>
- To: Scott McMorrow <scott@xxxxxxxxxxxxx>
- Date: Thu, 6 Dec 2007 13:00:00 -0800 (PST)
Scott,
I understood long ago that Yuriy is your friend. This is the reason why you
sent me his biography off-list. Since I didn't need it, I sent it back to you.
You don't need to mention what you did together 6 years ago. No body is
interested in this. We need explanation of the facts in this list, not personal
relationships.
I have said it openly in this list that my intention is not to attack anybody
personally. I don't know anything about Yuriy. So why should I attack him? Go
back and read all my mails. My aim was to clearify the facts on via-hole
modeling that can mislead people in this list. I'm happy you also acknowledge
the fact that his models were not fine. So what is your problem?
Go back and read Dr. Johnson's notes again on via-hole modeling. He treats
more than just the first appoximation you assume. It is an excellent text and
gives you a feeling of the relationship between signal and stitching vias. Not
just full wave simulations.
I don't know why you mentioned the propagation of higer order modes. No body
talked of that here. The discussion has been only on the case where the modes
are evanescent.
I can see how you get blind with friendship and can't even see the facts
anylonger.
Scott McMorrow <scott@xxxxxxxxxxxxx> wrote:
Charles,
You said:
" if you place your stitching vias (or whatever you call them), close to the
signal via, they will also define the inductance and impedance of the return
path. The inductance of the complete arrangment (signal via, stiching
vias/return path) depends almost entirely on the geometry of the vias and the
distance between the stitching vias and signal via. If you increase your
simulation area without increasing the geometry/position of the stitching vias,
then the inductance/impedance of your return path will not change."
This is only partially true, but it is not the complete story. Dr Johnson is
only partially correct on his treatment of via inductance. It is good as a 1st
approximation, but it does not tell the whole story. His treatment assumes
planar cavities of infinite extent and no full-wave effects occurring between
the signal, stitch vias, and planes. Planes are not infinite, and full-wave
effects do occur, however, under some conditions they can be approximated by
the quaint TEM phenomena that we call "inductance". Any time vias penetrate a
planar cavity, a circular parallel plate excitation will occur. Vias used to
stitch the cavity cannot contain all of the magnetic field and thus some energy
propagates into the cavity, depending on the frequency. The percentage of
energy lost to the cavity is frequency dependent, and will resonate and
eventually be converted to crosstalk, power system noise, and distortions in
the signal as the energy is re injected in the signal via. These
effects are non-localizable without considering the complete planar boundary
conditions, and constitute errors in modeling that are dependent on the exact
configuration of the planes. Many times these errors are acceptable. Some times
they are not. Frequency of operation, desired bandwidth, and required isolation
are determining factors in the acceptability of a particular modeling approach.
Placement of model launch ports at well-defined signal mode boundaries helps to
solve some of the localization problems, but not all of them. Signals following
vias go through very tricky rabbit holes indeed, and as soon as via lengths
approach significant fractions of the signal wavelength, large amounts of
energy can be converted to parallel plate mode excitation, well before other
higher order modes of propagation develop.
Yuriy and I have had many discussions about these things going back 5 or 6
years now. I'm quite sure he has an ample grasp on the RF Electromagnetic
phenomenon. I've also looked at his application notes and find that the are
fine for what they are, but in some cases they are quick and dirty. There's
nothing wrong with that! They serve to show and establish some fundamental
points. Knowing Yuriy, I know that he understands the inherent limitations of
the way that he launched into some of the structures, and I find his emails
quite enlightening as he expands on his thoughts.
Regards,
Scott
Scott McMorrow
Teraspeed Consulting Group LLC
121 North River Drive
Narragansett, RI 02882
(401) 284-1827 Business
(401) 284-1840 Fax
http://www.teraspeed.com
Teraspeed® is the registered service mark of
Teraspeed Consulting Group LLC
Charles Harrington wrote:
> Yuriy,
>
> if you place your stitching vias (or whatever you call them), close to the
> signal via, they will also define the inductance and impedance of the return
> path. The inductance of the complete arrangment (signal via, stiching
> vias/return path) depends almost entirely on the geometry of the vias and the
> distance between the stitching vias and signal via. If you increase your
> simulation area without increasing the geometry/position of the stitching
> vias, then the inductance/impedance of your return path will not change. So
> your explanation that changing the simulation area automatically changes the
> impedance of the return path and |S11|, is not correct. If you do not believe
> me, then contact Dr. Howard Johnson for further explanations. In his book
> (advanced black magic) he has an excellent explanation of the relationship
> between these stitching vias and the inductance of the return current of the
> signal via. This is the only text I know that treats these issues in a very
> practical manner and I think it will be of great help to you. I strongly
> recommend it to every other person working with via modeling. Note that the
> number of stitching vias you use depends on a number of other factors. From
> his book, you'll see just a method. You can apply the method to your own
> designs. Don't expect an automatic solution to your problem!
> Remember, Maxwell never said Ampere is wrong. He said Ampere's circuital law
> is not complete and then completed it using the displacement current term
> (See J.C. Maxwell - On Physical Lines of Force, 1861). Now, when your
> stitching vias define the return current path, the conduction current is much
> more greater than the inter-plane displacement currents, such that we can use
> the original form of Ampere's law without any loss of accuracy. Quasi-static
> methods use this approximation too. It's not that the displacement current is
> not present. But it is too small to be considered. With that said, you can
> visualize the discontinuity (considering its return paths and traces) as
> consisting of 2 parts. The closest part (which you beautifully describe as
> the "near field zone") and the "far-field zone" - if you allow me to use your
> words. In the "near field zone", which should be less than 3mm for
> frequencies up to 50 GHz (from my experience) when using typical PCB/package
> geometries, |S11| (at a given constant frequency) depends almost completely
> on the higher order modes excited by the discontinuity. |S11| decreases as
> the length increases because the higher order modes die off. In the far-field
> zone, |S11| increases because the higher order modes have compeltely decayed
> and other factors now play the deciding role. As you know, |S11| for a given
> line length at a givn frequency is constant.
> This is the method I have been using to define the boundaries of both
> vertical discontinuties(such as via-holes) and horizontal discontinuties
> (such as bends). And I'm fine with it. I have done many simulations and
> measurements. As I said earlier, I got it from the paper I cited on Nov. 20.
> and other publications from these authors. I had problems to accurately model
> discontinuties at higher frequencies (up to 77 GHz) years ago. Remember it is
> just a method and you have to take into consideration your real design
> enviroment, if you apply the method proposed in the paper. Don't expect an
> automatic solution to your problem! As far I know, the term "boundaries of
> discontinuities" as used in PCB/package design was first introduced by these
> authors. If you have your own method, then you can also use it. But one thing
> is clear. You have to define these boundaries at your 20 GHz and beyond. If
> not, your models are not correct and can not be validated with measurements.
> I'm not
> interested in your tool. Just like Chris, I'm interested in methods or
> methodology as he calls it.
>
> Hope this helps
>
> Best regards
> Charles
>
> Yuriy Shlepnev wrote:
> Chris,
>
> I think all definitions of the interconnect models you provided may be
> applicable under different circumstances. It depends on possibility to
> localize discontinuities such as vias and splits for the electromagnetic
> analysis. If all discontinuities in you channel can be isolated for the
> electromagnetic analysis, then the de-compositional analysis you described
> in a) and b) can be safely used. If such localization is impossible, the
> system-level model has to be built either on the base of a complete 3D
> electromagnetic analysis of the whole board (possible but not practical) or,
> alternatively, the de-compositional model has to be extended with models of
> such structures as parallel planes and splits with all decoupling structures
> connected to them (hybrid system-level models with transmission planes).
> Those structures are the major reasons of non-localizability of
> discontinuities on PCB and in packaging applications.
>
> How to define the boundaries of a discontinuity and localizability. If
> simulation results (S-parameters for instance) are relatively independent on
> the simulation area size and on the boundary conditions, the discontinuity
> can be isolated for the analysis and a reusable model can be generated
> (otherwise the discontinuity is not localizable). Simple rules based on line
> width/substrate height can be used to define the simulation area in case of
> localizable discontinuities. Phase reference planes may be shifted toward
> the discontinuity to make it electrically smaller (for better fitting or
> interpolation). A line segment with a minimal length to prohibit interaction
> through the higher order modes have to be added at the system-level in that
> case. This de-compositional technique used in microwave engineering since
> 40-s (Levin, Advanced theory of waveguides, 1951) and is the mainstream in
> the microwave system-level analysis tools. The smaller the localizable
> discontinuity the smaller the effective discontinuity area. It provides good
> models even for micron-sized structures up to sub-mm wave frequencies.
>
> Note that dependency of S-parameters on the simulation area sometime has
> nothing to do with the higher order modes discussed here before, but rather
> related to parallel planes and to localizability. S-parameters of a single
> via without or with a stitching via nearby can show significant dependency
> from the simulation area simply because of the discontinuity is not
> completely localized and the impedance of a cavity formed by parallel planes
> may change the |S11| for instance. Increasing the simulation area, one
> increase the inductance of the return path and together with the via
> capacitance to the planes it may be visible as the decrease of |S11| with
> the increase of the simulation area size (effect observed in the paper cited
> below). Such problems may be on the border line between the localizable and
> non-localizable problems and sometime may even require the system-level
> hybrid models with the transmission planes.
>
> Who has to define the boundaries of discontinuities or minimal length of the
> line segments to connect the discontinuities. Ideally, it has to be the
> system-level tool that decomposes a channel into transmission lines,
> discontinuities and possibly transmission planes. All mainstream SI tools
> are already based on the decomposition into line segments. It may include
> coupling between the lines bases on physical or electrical thresholds. The
> same approach has to be used to define what discontinuities may be analyzed
> with a 3D EM tool and what discontinuities require hybrid models. If two
> discontinuities are too close to each other (physical or electrical criteria
> can be used) - they have to be analyzed in a 3D solver as a whole and so on.
> In addition, a 3D solver has to define sufficient simulation area
> automatically and produce the model that is electrically as small as
> possible. Without such interaction between the system-level tool and a 3D
> solver you have to follow the recommendations provided by a 3D tool vendor
> and make sure that the discontinuity model is connected in the final design
> with sufficient line segments. I think that report on the minimal length of
> the line segments would be a good feature for an electromagnetic tool.
>
> Best regards,
> Yuriy
>
> Yuriy Shlepnev,
> Simberian Inc.
> www.simberian.com
>
> -----Original Message-----
> From: Chris Cheng [mailto:Chris.Cheng@xxxxxxxx]
> Sent: Thursday, November 29, 2007 6:18 PM
> To: ch_harrington@xxxxxxxxx; shlepnev@xxxxxxxxxxxxx
> Cc: SI LIST
> Subject: RE: [SI-LIST] Re: Signal crossing Split plane
>
> Charles and Yuriy,
> I have a philosophical question about modeling these 3D structures.
> It seems both of you agree that the entry ports needs to be back out to
> certain distant from the structure itself (most likely dimensionally
> compatible to the structure itself). So what is the definition of the
> overall system level interconnect model ?
> One can have the following defintions :
> a) interconnect model (most likely lossy trace model) with length up to the
> extended port location + the 3-D model of the plane cut/via transition model
> b) interconnect model (most likely lossy trace model) as report by the
> design data base + the 3-D model of the plane cut/via transition model -
> effect of just the extend port length of the interconnect model
> c) the entire interconnect structure is simulation in one gigantic 3D
> structure
>
> The combine last two terms of b) is what Roger Harrington used to call
> excess parasitics.
> In a PCB interconnect environment, a) and b) for all practical purpose are
> the same because the interconnect length >> extended port length. But for
> package model where the entire structures are measured in mm or mils, a) and
> b)
> may have significant differences.
>
> Should a 3D cad tool report the "excess parasitics" so that users can simply
> use the length report of the design database, then add in the via/plan cut
> section anytime he/she encounters such structure ?
> Or should a 3D cad tool be just modeling the true 3D structure but then has
> to warn users to back out the interconnect trace length to account for the
> extend port length (which seems to require careful consideration of the 3D
> structure on a case by case basis).
> Or, just lump the 3D structure into a gigantic 3D file together with the
> rest of the interconnect and pray that the simulator will converge ?
>
>
> -----Original Message-----
> From: si-list-bounce@xxxxxxxxxxxxx
> [mailto:si-list-bounce@xxxxxxxxxxxxx]On Behalf Of Charles Harrington
> Sent: Wednesday, November 21, 2007 2:57 PM
> To: shlepnev@xxxxxxxxxxxxx
> Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST'
> Subject: [SI-LIST] Re: Signal crossing Split plane
>
>
>
> Yuriy,
> I think we really have to end the discussion. I recommend you also
> talk to some experts in this forum about your models. They will tell you
> exactly what Im trying to say and even more.
> I didnt even know you have your own software. But I cannot understand why
> you make such claims that your software can compute whatever multilayered
> geometry and that it also automatically defines the boundary of the
> discontinuities. You know this is not true. We all know this is not true.
> So why do you make such claims? If I ask you, with what degree of accuracy
> does your software compute "whatever multilayer geometry" (when compared
> to measurements) and how does it automatically define the boundaries of
> discontinuities, I know that you will be baffled. So, I dont need the
> answers to these questions. However, I'm glad you acknowledge the fact
> that you need about 1mm distance away from the via pad at such higher
> frequencies to get accurate results. Let us leave it there. I will not
> write any more.
> I wish you the best with your models. 25 yrs of experience is quite a lot..
> I respect that. But as you can see, there is still a lot out there to
> learn.
> Best regards
> Charles
>
>
>
> Yuriy Shlepnev wrote: Hi Charles,
>
> Thank you for the reference. I am familiar with this paper as well as with
> the other publications of this group from Fraunhofer Institute.
> First of all, our 3D full-wave solver allows to build different via-hole
> models. It solves whatever multilayered geometry with ports you put in
> there.
> Second, the solver automatically defines the boundary of the
> discontinuities. See for instance the final model for optimal via-hole on
> slide 8 in
> http://www.simberian.com/Papers/OptimalDifViaholesDesign6pPCB.pdf. The
> differential line segment length in that particular example is about 1 mm,
> that is sufficient for high-order modes to die even at 30 GHz. Though, to
> define the area we use technique different from one described in the paper
> (I hinted details earlier). Lumped ports are often used for the
> preliminary
> optimization of via-holes because of it is quick and it provides good
> approximation (see for instance the final model and comments in the
> presentation mentioned above). Essentially, it ends the discussion.
> If you looked through the app notes on our web page you just saw the tip
> of
> an iceberg. We put about 25 years of research to develop and validate the
> technology. It is well documented on our web site in Downloads/Papers and
> Presentations areas.
> And, I do not even want to start discuss the definition of ports or
> multimodal decomposition, because of it looks strange to me that after
> reading Collins you still do not understand what it means and how it
> applies
> to the multilayered circuits.
>
> Best regards,
> Yuriy
>
> Yuriy Shlepnev, Ph.D.
> President, Simberian Inc.
> 2326 E Denny Way, Seattle, WA 98122, USA
> Tel/fax +1-206-726-1098
> Cell +1-206-409-2368
> Skype shlepnev
>
> www.simberian.com
>
>
>
>
>
> From: Charles Harrington [mailto:ch_harrington@xxxxxxxxx]
> Sent: Tuesday, November 20, 2007 2:46 PM
> To: shlepnev@xxxxxxxxxxxxx; scott@xxxxxxxxxxxxx
> Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST'
> Subject: RE: [SI-LIST] Re: Signal crossing Split plane
>
> Yuriy,
> I agree with some of your views. However, they contradict your via models..
> I couldn't reply yesterday, because I was trying search for the reference
> I
> mentioned, since you needed it. Many other people replied off-line and so
> needed the reference. Got it from IEEE Xplore.
>
> A Novel Methodology for Defining the Boundaries of Geometrical
> Discontinuities in Electronic Packages
> Ndip, I.; Reichl, H.; Guttowski, S.;
> Research in Microelectronics and Electronics 2006, Ph. D.
> 12- 15 June 2006 Page(s):193 - 196
>
> You mentioned in your mail that the near field zone as a result of the
> higher-order modes excited at the via expands with frequency and is very
> small. I agree with you.
> But the question is this. How small is it? How small or big is at 1 GHz,
> 10
> GHz, 20 GHz? Have you ever studied it? You have to take this zone into
> consideration when studying vias or any other structures that excite
> higher
> order modes.
> The method proposed in this paper is quite illustrative and useful. I
> understand it this way (Please correct me if I understand it wrongly):
> These higher-order modes (e.g., TE, TM...) are characteristics of the
> trace
> or transmission line and they die exponentially away from the point of
> excitation, i.e., the via-trace interface. S-parameters, like other
> network
> parameters, give us the relation between input and output signals. Now, to
> obtain S11, for example, you need to get the ratio of the reflected and
> input signals. Both signals must be of the same "type". We can not
> directly
> compare cars and aeroplanes, though both are used for transportation. You
> know your input signal (e.g., a transverse electromagnetic wave), because
> you excited it at the port. At discontinuities, an infinite order of
> given
> higher-order modes can be excited. The orders or strength of the excited
> modes differ from one discontinuity to another, although the modes can be
> the same. So, there is no way you can know all the orders of the
> higher-order modes excited and how they interact. Now if you place your
> ports quite close to the point of excitation of these modes, then your
> S-parameters must be wrong. Why? In this case, to obtain S11, you need to
> obtain the ratio of the unknown higher-order modes and your known excited
> transverse electromagnetic wave at the port. That's why in most 3D
> full-wave
> solvers, it is recommended that ports should be placed far away from the
> discontinuities, so as to enable these higher-order modes to die. When
> they
> die, then you can easily define your S-parameters which will then be the
> ratio of the input signal you know (transverse electromagnetic wave) and
> the
> reflected signal you know (transverse electromagnetic wave). To define the
=== message truncated ===
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