[SI-LIST] Re: Signal crossing Split plane
- From: Charles Harrington <ch_harrington@xxxxxxxxx>
- To: shlepnev@xxxxxxxxxxxxx, 'Chris Cheng' <Chris.Cheng@xxxxxxxx>
- Date: Mon, 3 Dec 2007 09:49:32 -0800 (PST)
Yuriy,
if you place your stitching vias (or whatever you call them), close to the
signal via, they will also define the inductance and impedance of the return
path. The inductance of the complete arrangment (signal via, stiching
vias/return path) depends almost entirely on the geometry of the vias and the
distance between the stitching vias and signal via. If you increase your
simulation area without increasing the geometry/position of the stitching vias,
then the inductance/impedance of your return path will not change. So your
explanation that changing the simulation area automatically changes the
impedance of the return path and |S11|, is not correct. If you do not believe
me, then contact Dr. Howard Johnson for further explanations. In his book
(advanced black magic) he has an excellent explanation of the relationship
between these stitching vias and the inductance of the return current of the
signal via. This is the only text I know that treats these issues in a very
practical manner and I think it will be of great help to you. I strongly
recommend it to every other person working with via modeling. Note that the
number of stitching vias you use depends on a number of other factors. From his
book, you'll see just a method. You can apply the method to your own designs.
Don't expect an automatic solution to your problem!
Remember, Maxwell never said Ampere is wrong. He said Ampere's circuital law
is not complete and then completed it using the displacement current term (See
J.C. Maxwell - On Physical Lines of Force, 1861). Now, when your stitching vias
define the return current path, the conduction current is much more greater
than the inter-plane displacement currents, such that we can use the original
form of Ampere's law without any loss of accuracy. Quasi-static methods use
this approximation too. It's not that the displacement current is not present.
But it is too small to be considered. With that said, you can visualize the
discontinuity (considering its return paths and traces) as consisting of 2
parts. The closest part (which you beautifully describe as the "near field
zone") and the "far-field zone" - if you allow me to use your words. In the
"near field zone", which should be less than 3mm for frequencies up to 50 GHz
(from my experience) when using typical PCB/package
geometries, |S11| (at a given constant frequency) depends almost completely on
the higher order modes excited by the discontinuity. |S11| decreases as the
length increases because the higher order modes die off. In the far-field zone,
|S11| increases because the higher order modes have compeltely decayed and
other factors now play the deciding role. As you know, |S11| for a given line
length at a givn frequency is constant.
This is the method I have been using to define the boundaries of both
vertical discontinuties(such as via-holes) and horizontal discontinuties (such
as bends). And I'm fine with it. I have done many simulations and measurements.
As I said earlier, I got it from the paper I cited on Nov. 20. and other
publications from these authors. I had problems to accurately model
discontinuties at higher frequencies (up to 77 GHz) years ago. Remember it is
just a method and you have to take into consideration your real design
enviroment, if you apply the method proposed in the paper. Don't expect an
automatic solution to your problem! As far I know, the term "boundaries of
discontinuities" as used in PCB/package design was first introduced by these
authors. If you have your own method, then you can also use it. But one thing
is clear. You have to define these boundaries at your 20 GHz and beyond. If
not, your models are not correct and can not be validated with measurements.
I'm not
interested in your tool. Just like Chris, I'm interested in methods or
methodology as he calls it.
Hope this helps
Best regards
Charles
Yuriy Shlepnev <shlepnev@xxxxxxxxxxxxx> wrote:
Chris,
I think all definitions of the interconnect models you provided may be
applicable under different circumstances. It depends on possibility to
localize discontinuities such as vias and splits for the electromagnetic
analysis. If all discontinuities in you channel can be isolated for the
electromagnetic analysis, then the de-compositional analysis you described
in a) and b) can be safely used. If such localization is impossible, the
system-level model has to be built either on the base of a complete 3D
electromagnetic analysis of the whole board (possible but not practical) or,
alternatively, the de-compositional model has to be extended with models of
such structures as parallel planes and splits with all decoupling structures
connected to them (hybrid system-level models with transmission planes).
Those structures are the major reasons of non-localizability of
discontinuities on PCB and in packaging applications.
How to define the boundaries of a discontinuity and localizability. If
simulation results (S-parameters for instance) are relatively independent on
the simulation area size and on the boundary conditions, the discontinuity
can be isolated for the analysis and a reusable model can be generated
(otherwise the discontinuity is not localizable). Simple rules based on line
width/substrate height can be used to define the simulation area in case of
localizable discontinuities. Phase reference planes may be shifted toward
the discontinuity to make it electrically smaller (for better fitting or
interpolation). A line segment with a minimal length to prohibit interaction
through the higher order modes have to be added at the system-level in that
case. This de-compositional technique used in microwave engineering since
40-s (Levin, Advanced theory of waveguides, 1951) and is the mainstream in
the microwave system-level analysis tools. The smaller the localizable
discontinuity the smaller the effective discontinuity area. It provides good
models even for micron-sized structures up to sub-mm wave frequencies.
Note that dependency of S-parameters on the simulation area sometime has
nothing to do with the higher order modes discussed here before, but rather
related to parallel planes and to localizability. S-parameters of a single
via without or with a stitching via nearby can show significant dependency
from the simulation area simply because of the discontinuity is not
completely localized and the impedance of a cavity formed by parallel planes
may change the |S11| for instance. Increasing the simulation area, one
increase the inductance of the return path and together with the via
capacitance to the planes it may be visible as the decrease of |S11| with
the increase of the simulation area size (effect observed in the paper cited
below). Such problems may be on the border line between the localizable and
non-localizable problems and sometime may even require the system-level
hybrid models with the transmission planes.
Who has to define the boundaries of discontinuities or minimal length of the
line segments to connect the discontinuities. Ideally, it has to be the
system-level tool that decomposes a channel into transmission lines,
discontinuities and possibly transmission planes. All mainstream SI tools
are already based on the decomposition into line segments. It may include
coupling between the lines bases on physical or electrical thresholds. The
same approach has to be used to define what discontinuities may be analyzed
with a 3D EM tool and what discontinuities require hybrid models. If two
discontinuities are too close to each other (physical or electrical criteria
can be used) - they have to be analyzed in a 3D solver as a whole and so on.
In addition, a 3D solver has to define sufficient simulation area
automatically and produce the model that is electrically as small as
possible. Without such interaction between the system-level tool and a 3D
solver you have to follow the recommendations provided by a 3D tool vendor
and make sure that the discontinuity model is connected in the final design
with sufficient line segments. I think that report on the minimal length of
the line segments would be a good feature for an electromagnetic tool.
Best regards,
Yuriy
Yuriy Shlepnev,
Simberian Inc.
www.simberian.com
-----Original Message-----
From: Chris Cheng [mailto:Chris.Cheng@xxxxxxxx]
Sent: Thursday, November 29, 2007 6:18 PM
To: ch_harrington@xxxxxxxxx; shlepnev@xxxxxxxxxxxxx
Cc: SI LIST
Subject: RE: [SI-LIST] Re: Signal crossing Split plane
Charles and Yuriy,
I have a philosophical question about modeling these 3D structures.
It seems both of you agree that the entry ports needs to be back out to
certain distant from the structure itself (most likely dimensionally
compatible to the structure itself). So what is the definition of the
overall system level interconnect model ?
One can have the following defintions :
a) interconnect model (most likely lossy trace model) with length up to the
extended port location + the 3-D model of the plane cut/via transition model
b) interconnect model (most likely lossy trace model) as report by the
design data base + the 3-D model of the plane cut/via transition model -
effect of just the extend port length of the interconnect model
c) the entire interconnect structure is simulation in one gigantic 3D
structure
The combine last two terms of b) is what Roger Harrington used to call
excess parasitics.
In a PCB interconnect environment, a) and b) for all practical purpose are
the same because the interconnect length >> extended port length. But for
package model where the entire structures are measured in mm or mils, a) and
b)
may have significant differences.
Should a 3D cad tool report the "excess parasitics" so that users can simply
use the length report of the design database, then add in the via/plan cut
section anytime he/she encounters such structure ?
Or should a 3D cad tool be just modeling the true 3D structure but then has
to warn users to back out the interconnect trace length to account for the
extend port length (which seems to require careful consideration of the 3D
structure on a case by case basis).
Or, just lump the 3D structure into a gigantic 3D file together with the
rest of the interconnect and pray that the simulator will converge ?
-----Original Message-----
From: si-list-bounce@xxxxxxxxxxxxx
[mailto:si-list-bounce@xxxxxxxxxxxxx]On Behalf Of Charles Harrington
Sent: Wednesday, November 21, 2007 2:57 PM
To: shlepnev@xxxxxxxxxxxxx
Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST'
Subject: [SI-LIST] Re: Signal crossing Split plane
Yuriy,
I think we really have to end the discussion. I recommend you also
talk to some experts in this forum about your models. They will tell you
exactly what Im trying to say and even more.
I didnt even know you have your own software. But I cannot understand why
you make such claims that your software can compute whatever multilayered
geometry and that it also automatically defines the boundary of the
discontinuities. You know this is not true. We all know this is not true.
So why do you make such claims? If I ask you, with what degree of accuracy
does your software compute "whatever multilayer geometry" (when compared
to measurements) and how does it automatically define the boundaries of
discontinuities, I know that you will be baffled. So, I dont need the
answers to these questions. However, I'm glad you acknowledge the fact
that you need about 1mm distance away from the via pad at such higher
frequencies to get accurate results. Let us leave it there. I will not
write any more.
I wish you the best with your models. 25 yrs of experience is quite a lot.
I respect that. But as you can see, there is still a lot out there to
learn.
Best regards
Charles
Yuriy Shlepnev wrote: Hi Charles,
Thank you for the reference. I am familiar with this paper as well as with
the other publications of this group from Fraunhofer Institute.
First of all, our 3D full-wave solver allows to build different via-hole
models. It solves whatever multilayered geometry with ports you put in
there.
Second, the solver automatically defines the boundary of the
discontinuities. See for instance the final model for optimal via-hole on
slide 8 in
http://www.simberian.com/Papers/OptimalDifViaholesDesign6pPCB.pdf. The
differential line segment length in that particular example is about 1 mm,
that is sufficient for high-order modes to die even at 30 GHz. Though, to
define the area we use technique different from one described in the paper
(I hinted details earlier). Lumped ports are often used for the
preliminary
optimization of via-holes because of it is quick and it provides good
approximation (see for instance the final model and comments in the
presentation mentioned above). Essentially, it ends the discussion.
If you looked through the app notes on our web page you just saw the tip
of
an iceberg. We put about 25 years of research to develop and validate the
technology. It is well documented on our web site in Downloads/Papers and
Presentations areas.
And, I do not even want to start discuss the definition of ports or
multimodal decomposition, because of it looks strange to me that after
reading Collins you still do not understand what it means and how it
applies
to the multilayered circuits.
Best regards,
Yuriy
Yuriy Shlepnev, Ph.D.
President, Simberian Inc.
2326 E Denny Way, Seattle, WA 98122, USA
Tel/fax +1-206-726-1098
Cell +1-206-409-2368
Skype shlepnev
www.simberian.com
From: Charles Harrington [mailto:ch_harrington@xxxxxxxxx]
Sent: Tuesday, November 20, 2007 2:46 PM
To: shlepnev@xxxxxxxxxxxxx; scott@xxxxxxxxxxxxx
Cc: sunil_bharadwaz@xxxxxxxxx; 'SI LIST'
Subject: RE: [SI-LIST] Re: Signal crossing Split plane
Yuriy,
I agree with some of your views. However, they contradict your via models.
I couldn't reply yesterday, because I was trying search for the reference
I
mentioned, since you needed it. Many other people replied off-line and so
needed the reference. Got it from IEEE Xplore.
A Novel Methodology for Defining the Boundaries of Geometrical
Discontinuities in Electronic Packages
Ndip, I.; Reichl, H.; Guttowski, S.;
Research in Microelectronics and Electronics 2006, Ph. D.
12- 15 June 2006 Page(s):193 - 196
You mentioned in your mail that the near field zone as a result of the
higher-order modes excited at the via expands with frequency and is very
small. I agree with you.
But the question is this. How small is it? How small or big is at 1 GHz,
10
GHz, 20 GHz? Have you ever studied it? You have to take this zone into
consideration when studying vias or any other structures that excite
higher
order modes.
The method proposed in this paper is quite illustrative and useful. I
understand it this way (Please correct me if I understand it wrongly):
These higher-order modes (e.g., TE, TM...) are characteristics of the
trace
or transmission line and they die exponentially away from the point of
excitation, i.e., the via-trace interface. S-parameters, like other
network
parameters, give us the relation between input and output signals. Now, to
obtain S11, for example, you need to get the ratio of the reflected and
input signals. Both signals must be of the same "type". We can not
directly
compare cars and aeroplanes, though both are used for transportation. You
know your input signal (e.g., a transverse electromagnetic wave), because
you excited it at the port. At discontinuities, an infinite order of
given
higher-order modes can be excited. The orders or strength of the excited
modes differ from one discontinuity to another, although the modes can be
the same. So, there is no way you can know all the orders of the
higher-order modes excited and how they interact. Now if you place your
ports quite close to the point of excitation of these modes, then your
S-parameters must be wrong. Why? In this case, to obtain S11, you need to
obtain the ratio of the unknown higher-order modes and your known excited
transverse electromagnetic wave at the port. That's why in most 3D
full-wave
solvers, it is recommended that ports should be placed far away from the
discontinuities, so as to enable these higher-order modes to die. When
they
die, then you can easily define your S-parameters which will then be the
ratio of the input signal you know (transverse electromagnetic wave) and
the
reflected signal you know (transverse electromagnetic wave). To define the
points where these modes die or have attenuated substantially, these
authors
argued that near the discontinuity, the imaginary part of the Poynting
vector describes the reactive energy associated with these higher-order
modes. So they studied this imaginary part and used it to define the point
where the modes die. I think they mentioned that only at a distance of
about
1mm away from the via-trace interface, at 20 GHz (or may be 30 GHz) may
you
place your ports, to get correct results. Certainly, this depends on the
via
geometry and trace type. But I find the results very helpful and can be
used
as a base for further experiments. You can get the details from the paper.
Unfortunately in your case, you compare what you don't know (reflected
signal) and what you know (excited input signal). In your via models,
neither did you define the required distance away from the via-trace
interface needed for these modes to die nor did you follow the advice
given
in full-wave solvers to be far way from the via-trace interface. You
considered the via just as the barrel and the pads at 20 GHz and beyond.
That's why I mentioned yesterday that your via models are not correct and
your S-parameter results are misleading. If you wish to study only the
behaivor of the barrel alone at lower frequencies (for what ever reason -
but not for realistic designs), then you don't even need a field solver.
You
can get formulas from good SI texts like that of Horward Johnson or from
papers.
At first I was also making the same mistakes as you are making right now.
I
had a lot of difficulties to correlate my simulation and measurement
results. So I learnt a lot from this paper, from Professor C. Balanis
(Advanced engineering electromagnetics) and from Professor R. Collins
(Field
theory of guided waves). I think these references will be good for you.
You
need all three of them.
There are also a lot of points that you need to modify in your models.
It's ridiculous when you talk of -30 dB attenuation of higher-order modes.
Which higher-order mode? Which order of this mode? Basic electromagnetic
theory teaches us that an infinite order of a given higher-order mode can
be
excited at any discontinuity. An interaction between makes matters worst.
So
how do you separate the different orders of the modes and tell which one
attenuates by -30 dB? Are the modes propagating or evanescent? Never use
rule of thumbs that have no base. I supposed you meant attenuation of the
fundamental mode which is propagating.
I don't know anything about the lumped ports you use. All I know is that
some lumped ports in some field solvers assume perfect H boundary
conditions
on the sides. Consequently, depending you may not even capture stray
fields.
So you can even get the worst results with lumped ports.
You can only shift your reference S-parameters plane and get accurate
results if your model captured all the necessary field behavior. But you
can
not simulate the via and traces differently and then do some
post-processing
or circuit modeling afterwards and expect to get correct results at higher
frequencies. The traces too are part of the "via effect" at least, at the
frequencies you are interested in (20 GHz and beyond), because the stored
higher-order modes give rise to additional inductances and capacitances.
These inductances and capacitances can not be captured if you analyze the
vias separately from their traces.
Finally, the theory of multi-modal decomposition means different things to
different electrical engineers. So I don't know what you mean. If you mean
that different parts of a system can be analyzed separately and then put
together, then it's true that it has been done for decades now. But the
question is this. How do you bring the different parts together in the
case
where there are discontinuities like vias? How do you define the via? How
small or big is your near field zone? I bet you, we have not yet
understood
this type of decomposition and it has not been done, or at least published
for decades. Whenever we have to deal with vias and other discontinuities
at
higher frequencies, straight-forward modeling can not be used.
Please Yuryi, don't get me wrong. I'm not trying to highlight on your
errors. I have mine too, like any body else. No one is perfect. I'm just
trying to raise the point that we need to be careful when modeling vias at
your frequencies. I agree with most of the points you made, but disagree
on
the ones stated above. We learn from each other when we exchange ideas
about
such fundamental issues that affect our modeling results. I think that is
one of the reasons why Ray and his team set up this forum.
Best regards.
Charles
Yuriy Shlepnev wrote:
Charles,
I am sorry that the simulation examples were not helpful to you. I will
appreciate if you send me the reference you mentioned - I am preparing to
be
shocked:)
You are absolutely right, the via-holes are not just pads and barrels and
there is no one solution that covers all possible cases. Analysis of
different vias has to be done in different ways. Transition to the traces
have to be almost always included in the final model for analysis of
multi-gigabit channels. Moreover sometime the via-hole problem cannot be
solved locally and require analysis of parallel plane structures with all
decoupling structures attached (see technical presentation #1 at
http://www.simberian.com/Presentations.php for more details on different
structures).
Considering the ports and excitation. Analysis of via-holes with lumped
ports provides just rough idea about the via-hole behavior. It is similar
to
what you would see from a differential probe attached to the pads of the
via-holes. Transition to traces and transmission line or wave-ports have
to
be used for the final extraction of S-parameters for the system-level
analysis (I am sorry that you missed this part in app notes). Note that it
is possible only for the localizable via-holes or via-holes not coupled to
parallel planes in general. Such t-line ports have to be positioned at a
distance from the via-hole that guaranties that the high-order modes are
attenuated substantially (for practical applications we usually use -30 dB
threshold at the highest frequency of interest). After such analysis, the
phase reference planes of S-parameters can be safely shifted closer to the
via-hole at the position where t-lines are still continuous to preserve
causality (to the edges of anti-pads for instance). Such transformation
does
not affect the near field or high order modes around the via-holes and the
final model can be safely connected with the transmission line segments in
a
system-level solver. Though, the model have to be used with transmission
line segments with length not less than in the electromagnetic analysis
(to
avoid the near-field interaction between the vias and possible
discontinuities). This technique called the multi-modal de-compositional
analysis and used in microwave engineering for decades at frequencies even
higher than 20 GHz.
Note, that in typical PCB trace the cut-off frequencies for high-order
modes
are extremely high. 10 mil trace on 10 mil dielectric with dielectric
constant 4.2 have cut-off frequency about 120 GHz, and the cross-over with
the surface TM mode may happen only at 200 GHz. Before 120 GHz the
high-order modes are evanescent and essentially form the via-hole near
field. This near-field zone is expanding with the frequency, but at 20 GHz
the area is still relatively small. Thus S-parameters only for the
dominant
modes can be safely extracted and used as the via-hole model.
Cases when via-hole excite the non-evanescent parallel-plane modes and
planes are not stitched close to the via-hole cannot be solved locally
=== message truncated ===
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